Controller

ABSTRACT

An electrical controller for electric motors is provided. A control system for an electric motor comprises a supply for supplying excitation current to different windings of the motor at any given time. Furthermore, the amplitude of the excitation current is independently variable of the timing and duration of the application of the excitation current to the windings. This allows increased control of the motor and facilitates the operation of the motor at high mechanical and/or electrical speeds.

This present invention relates to an electrical controller for electricmotors. In particular, the invention relates to a system for improvingthe control and operation of Alternating Current motors, whereinpermanent magnets or steadily excited electromagnets move in thepresence of stationary electrical windings. These motors will bereferred to collectively as permanent magnet Alternating Current (PMAC)motors (although some may substitute steadily excited electromagnets forpermanent magnets). PMAC motors may be found, for example, in high-speedhybrid turbochargers or other high-speed electrical devices.

Conventional PMAC motors utilise the movement of permanent magnets inthe presence of stationary electrical windings. The stator windings mustbe excited by an oscillating or intermittent electrical current (i.e.,AC or PWM) in order to exert an electromotive force upon the magnets asthe magnets rotate or translate relative to the windings. Such motorsare typically described as brushless alternating current permanentmagnet motors or permanent magnet synchronous motors (PMSM). It isimportant to note that such motors are distinct from brushless directcurrent permanent magnet motors that have a different construction andcontrol methodology.

Brushless AC permanent magnet motors are among the most mechanicallysimple, compact, and efficient types of motors. However, throughout thehistory of electric motors, practical embodiments have usuallyincorporated design features which compromise simplicity, compactness,and efficiency in order to impart favourable operating characteristicsthat simplify the task of controlling the motor. Examples of compromisesinclude:

-   -   1. field weakening to limit speed from the inherent properties        of the motor;    -   2. helical magnets to improve starting torque and predictability        at low speeds;    -   3. electromagnets rather than permanent magnets to allow motor        torque to be adjusted by the direct current signal powering the        electromagnets;    -   4. a distribution of stator windings chosen in such a way as to        smooth the torque output of the motor given a smooth alternating        current (oscillating) input;    -   5. variable air gap (especially in ‘axial flux’ type motors) to        allow the motor constant (relationship between current input and        torque output) to be adjusted by mechanical means; and    -   6. the use of weaker magnets or passively excited (metallic)        materials to reduce the sensitivity of the motor to the shape of        the input alternating current signal.

Electric motors usually operate at speeds below 200 Hz (e.g., electriccars=20 Hz to 100 Hz, automotive starter motors=30 Hz to 50 Hz, UK powerstation generators=50 Hz, typical pump motors=50 Hz, domesticappliances=10 Hz to 50 Hz, conveyors and pulleys=1 Hz to 50 Hz).

High-speed applications have typically favoured design approach number 7from the above list: the use of passively excited materials. Examplesinclude: J R Bumby, E Spooner, & M Jagiela, “Solid Rotor InductionMachnies for use in Electrically-Assisted Turbochargers”, Proceedings ofthe XVII International Conference on Electric Machines (ICEM), 2006; andS Calverly, “High-speed switched reluctance machine for automotiveturbo-generators”, Mag. Soc. Seminar on Motors and Actuators forAutomotive Applications, 2002.

Incorporation of the above design features significantly adds to thesize, weight, cost and energy efficiency of motors. Additionally, forany chosen design, motor size, weight, and cost are typicallyproportional to torque output. Motors that operate at higher speeds candeliver equivalent power at lower torque, so equipment and transmissionsare often specified to accommodate higher-speed motors where possible.However, higher speeds tend to exacerbate the challenges associated withmotor control.

One application area of particular interest is turbomachinery. Thesedevices, which operate upon gas at speeds approaching the sound barrier,spin at speeds in excess of 1,500 Hz. Turbomachines are well-known inaerospace and natural gas power generation, but they are increasinglyfound in automotive engines (turbochargers), industrial processes(compressors and heat recovery systems), domestic appliances (vacuumcleaners), and building heating and ventilation. The increasingpopularity of turbo machines in comparison to fixed-displacement pumpsand expansion chambers has created a further demand for high-speedmotors and provides the prospect of reducing or eliminating the cost ofhigh-ratio transmissions if higher-speed motors can be supplied. Ofparticular interest for the present invention is the electrification ofthe automotive turbocharger, which is the subject of an earlier patent(B Richards, “Turbocharger concept”, UK Patent no. 0624599.7, 2006).

Turbomachines require operating speeds in excess of 1,500 Hz, and someautomotive applications require speeds above 2,500 Hz. Typical motorspeeds below 200 Hz are not suitable for this application. Designcompromises which use weaker magnets or passively magnetised materialscan achieve speeds approaching 1,500 Hz, but these have relatively lowpower density. A typical 20 kW turbo compressor is approximately 15cm×15 cm×10 cm in shape and requires approximately 1.6 Nm torque inputin steady state. A typical passively excited electric motor operating atthe same speed could supply the required torque from a sufficientlylarge motor. But such a motor would have a large rotor inertia. As themotor increases further in size to provide excess torque to overcome itsown inertia during transient acceleration, the inertia of the motorincreases in proportion to the extra torque produced, giving diminishingreturns. By contrast, a motor with strong permanent magnets can achieve10 times the torque from the same volume, allowing the motor to besmaller (on the order of 10 cm×10 cm×10 cm for 2 Nm) while stillproviding sufficient torque for acceleration. A problem of managingcontroller current remains.

Because of the design advantages described above and the emergingapplications for high-speed machines, there has been an overall trend inthe past three decades towards motors that are ever more difficult tocontrol. This trend has coincided with, and depended upon, thewidespread improvement of electronics and computers that enable evermore sophisticated control strategies.

Conventional brushless permanent magnet motors are generally either ofthe DC or AC type. Brushless DC motors accept ‘rough’ voltage input andsmooth the flow of current internally by the inductance and resistanceof the motor windings. Brushless AC motors (also called synchronousmotors) require a smooth, sinusoidal (or near sinusoidal) current to beimparted by the controller. Neither is designed to accept ‘rough’waveforms of current input.

The conventional approach to controlling a brushless permanent magnetmotor is pulse width modulation (PWM). An example of this approach(relating specifically to PMAC) is shown in EP 2,159,909. This documentutilises rapid PWM to simulate a smooth sine wave voltage input to themotor. This allows for precise control of position and smooth operationof the motor (especially at low speeds).

Brushless DC permanent magnet motors also use PWM to control theamplitude and phase of the motor's input voltage. The principledistinction between a brushless DC and a brushless AC permanent magnetmachine is that a brushless AC motor additionally requires its PWMcontroller to synthesize a sinusoidal signal, while DC allows the PWMoutput to be a ‘rough’ voltage waveform. In either case (DC or AC), PWMgenerates a motor control signal of fixed amplitude and frequency andapplies the signal at each commutation. The supply of the requiredsignal, for example ‘rough’ (DC) or sinusoidal (AC), is achieved byvarying the number and duration of pulses supplied to the motor. Thisgenerally entails providing several pulses per commutation of the motorto try to approximate the ideal waveform input required for the motorused. The overall amplitude (or voltage) supplied by PWM is thereforecontrolled by varying the number and duration of pulses supplied to themotor for any given phase.

However, use of a PWM controller is calculation intensive andnecessitates the controller to operate at a frequency at least 10x(typically 100× or more) in excess of the rotational frequency of themotor. This means, for example, that an automotive turbochargercompressor would require a controller with at least 15,000 Hz internaloperating frequency. This is well within reach for low-power logiccircuits, but it approaches the limits of what can be achieved withhigh-powered electronic circuitry today.

A controller that may be embodied by the invention of the presentapplication which will be described below produces ‘rough’ wave forms ofcurrent and requires (or corresponds to) a motor with an atypicaldesign.

The proposed motor is of the brushless permanent magnet type, withproperties that are different from either a typical brushless AC orbrushless DC permanent magnet motor. The properties of this motor areknown in the art, but this selection and combination of properties isunusual. Specifically, this motor has properties which enhance itsability to accept ‘rough’ waveforms of current (or, in the case of agenerator, its tendency to produce ‘rough’ waveforms) and may be used inan advantageous way with the controllers embodying the presentinvention. Its properties are:

-   -   a. rotor magnets made of materials with strong permanent magnet        properties and shaped in such a way as to have constant        thickness through the angular dimension corresponding to shaft        rotation and distributed about the rotor without gaps (all of        which design features cause the electromagnetic field        experienced by the windings near the magnets' edge to be similar        in strength to the electromagnetic field experienced near the        middle of the magnets);    -   b. a number of teeth (metal elements around which stator        windings are wrapped) which is divisible by the number of        electrical phase connections provided by the motor and a winding        pattern such that any collection of series- or        parallel-connected windings which constitutes a phase will        everywhere be subjected to an identical (varying through shaft        rotation but everywhere and at all times equivalent to each        other) electromagnetic field arising from the motor magnets (so        that all the windings in a phase are complementary and do not        serve to counteract one another, at any rotational angle of the        shaft);    -   c. a number of magnets (‘poles’) chosen such that b is        implementable;    -   d. magnet angular thickness (arc length) and winding pitch        (number of teeth spanned by a single winding loop) chosen so        that the angle of rotation of the rotor through which an        interface between adjacent magnets of opposing polarity passes        within the span of a winding loop matches precisely the phase        angle (proportion of the period of oscillation) through which        the controller will maintain maximum current through that        winding loop; and    -   e. minimum winding inductance achieved by a preference towards        the individual wire loops forming the windings being connected        in parallel rather than series to the extent allowed by other        considerations such as the current and voltage specifications to        which the motor must conform (in recognition of the fact that        greater internal inductance will tend to smooth and delay the        signals produced by the controller, with the extent of such        transformation being a function of motor speed and thus        difficult to accommodate within the controller).

These features, although individually known in an academic sense, arenot believed to be used in combination in typical, commercially-builtelectric motors, nor are the impact of these design features upon thecontroller widely considered or understood. In fact, the amalgamationwithin one organisation of both motor design and motor control isatypical within the industry. It will also be appreciated that thiscombination of features can be used to create a generator with uniquecharacteristics that impact upon the design of the generator controller.In particular, the generator will provide an output that lends itself toDC rectification.

It is an aim of the present invention to provide a controller that canprovide high electrical speed drive using power electronics. The controlapproach described here is substantially different from the conventionalpulse width modulation approach, which synthesises an entire waveformfrom one switching process. A pulse width modulation controller wouldrequire an output load with high inductance, in order to smooth thesuccessive pulses into an alternating waveform. A brushless DC motor,with a square-wave voltage input would likewise require high inductance.This requirement for high inductance can also be a limitation in theperformance of the motor, especially at high mechanical and electricalspeed.

According to the present invention, there is provided a control systemfor an alternating current electric motor comprising means for supplyingexcitation current to different windings of the motor at any given time,wherein the amplitude of the excitation current is independentlyvariable of the timing and duration of excitation current delivery.

In a preferred embodiment, the supply means comprises an excitationcurrent supply feedback loop for providing a current amplitude (whichcorresponds to an ‘aggregate current’) to the motor windings, and acommutation feedback loop separate from the excitation current supplyfeedback loop for controlling the timing and duration of excitationcurrent supplied to the motor windings. Said excitation current may bean aggregate current. The current supply feedback loop may comprise aninput for representing a target motor speed, and means for providing acurrent amplitude responsive to the target speed of the motor and themotor speed signal. The current supply feedback loop may furthercomprise a regulating feedback loop for regulating the currentamplitude.

Means, which may be in the form of a sensor, may be provided forproviding a signal indicative of the motor speed and/or the angularposition of the motor. The angular position indication may be a rough orquantized measurement, not necessarily a fine angular measurement. Thismeans may be a commutation signal or current indicative of the currentinduced in the motor windings.

The commutation feedback loop may be operative for the control of thetiming and duration of the current pulses to the motor windings independence upon the angular position signal. The movement of the motorgenerates a back EMF in each phase winding that corresponds to aposition signal. The commutation feedback loop provides a current pulseto a given phase winding when the amplitude of the sensed back EMF forthe corresponding phase winding is greater than the back EMF sensed inrespect of the other ones of the plurality of phase windings. Thecommutation feedback loop may comprise a filter for filtering the sensedback EMF to smooth the signal and/or to facilitate synchronisationbetween the supply of the current to said given winding and the angularposition signal. The filter may be operative to introduce a phase shiftto the sensed back EMF to facilitate said synchronisation. The motor maycomprise three phase windings at 120 degree relative angulardisplacement. Also to facilitate synchronisation between the supply ofcurrent and the angular position sensor, the back EMF used to sensemotor position may not be measured from individual phase windings butfrom combinations thereof (for example, the sum or difference of backEMF across two phase windings), which may entail a known phase offsetfrom the back EMF present on an individual phase

In an alternative embodiment, the sensor for providing a signalindicative of the motor speed and/or the angular position of the motormay be a Hall Effect sensor or an optical sensor associated with anoutput shaft of the motor.

According to the present invention there is further provided a controlsystem for an alternating current electric motor, the control systemcomprising a commutation circuit operative for controlling a timing andduration of current supplied to the electric motor, and a power supplyfor supplying current to the motor, wherein the commutation circuit isoperationally independent of the power supply.

According to the present invention there is yet further provided apermanent magnet motor comprising a control system as defined above,wherein the motor includes a plurality of permanent magnets disposedabout a rotor such as to provide a continuous magnetic shell withoutgaps between the magnets. It may be appreciated that the permanentmagnets utilised by such a motor can be any magnetised material wherethe control the motor does not rely upon the ability to vary themagnetic strength. The motor may comprise a plurality of windingsdisposed about the magnetic surface in a number of slots such that eachslot may be energised by a single current pulse.

Furthermore, an ideal arrangement for the permanent magnet motorprovides that the number of slots arranged around the circumference ofthe motor, divided by the number of magnets disposed about the rotor ofthe motor is a multiple of the number of phases of the current. It maybe appreciated that the number of windings of the motor may be tailoredto achieve this ratio.

According to the present invention there is also further provided aforced induction system for an internal combustion engine comprising acontrol system as described above. Such forced induction system may be asupercharger or may be a turbocharger. In an exemplary aspect of theinvention, the forced induction system is for an internal combustionengine, wherein the system comprises a compressor, a turbine, agenerator, an electric motor and a control system. In said aspect, thecompressor acts to increase the pressure of gas into the engine and ismechanically decoupled from the turbine which is arranged to be drivenby the engine exhaust gas and to drive a generator electricallyconnected to the electric motor. In turn, the compressor is driven bythe electric motor and therefore the compressor is driven, at least inpart, by the output torque of the turbine via the electrical connection.In such aspects, the electric motor is controlled by the control systemdescribed above.

In a further aspect of the present invention, there is provided agenerator for generating electric power, the generator comprising: arotor having a plurality of permanent magnets evenly and continuouslydistributed around the motor; a stator having a plurality of windings,wherein rotation of the rotor relative to the stator windings generate aplurality of phase shifted square waveforms; and a control circuit,wherein the control circuit comprises a commutation circuit arranged todraw current from the stator windings, said control circuit controllingthe timing and duration of drawing off of the current from the windingsindependently of the rotation of the rotor.

In an alternative or additional aspect, there is provided a permanentmagnet generator including a plurality of permanent magnets in contactwith one another without spaces between the magnets such as to provide acontinuous permanent magnet shell. Such a permanent magnet generator maycomprise a plurality of phase windings disposed in a number of slotsaround the circumference of the generator or permanent magnet shell suchthat each phase winding may be complementarily energised by a singleexcitation current and do not serve to counteract one another.Furthermore, said generator may be arranged such that the plurality ofphase windings are individually disposed within the slots in paralleland such that the number of slots divided by the number of magnets is amultiple of the number of phases of the excitation current.

Embodiments of the invention are advantageous in that they provide forvariation of the motor speed (torque) independently of the commutation.This is effected by supply of an aggregate current to the motorindependently to the commutation that directs that current to theappropriate motor phase winding(s). Aggregate current can be regulatedusing a PWM approach, and there may be a requirement for inductancebetween the current supply controller and the commutation controller inorder to smooth that PWM output. However, the frequency of the PWMcurrent supply controller and the inductance that may be required toregulate the output of the current supply controller are parameters thatare essentially independent from the inductance and rotational speed ofthe motor. This means that the frequency of the PWM signal need not behigher than the motor's operating frequency, as in prior artarrangements. The commutation controller, which sits between the currentsupply controller and the motor, does not implement PWM. Rather, thecommutation controller simply directs the current to the correct windingaccording to the angular position of the motor. Because the commutationcontroller does not implement PWM, its design is relatively simple andit can achieve high motor operating speeds. Also because the commutationcontroller does not implement PWM, the motor need not exhibit highinductance and its efficiency (especially at high electrical/mechanicalspeeds) is thus improved. This means that higher motor speeds arepossible without the need for one integrated high power electroniccontroller to provide for the commutation and also regulate totalcurrent in a single step.

The invention will now be further described by way of example withreference to the accompanying drawings, in which:

FIG. 1 a is a waveform diagram of a conventional motor;

FIG. 1 b represents the structure of the conventional motor of FIG. 1 a;

FIG. 1 c is a waveform diagram for an alternative conventional motor;

FIG. 2 a is a waveform diagram of a motor that can be utilised inembodiments of the present invention;

FIG. 2 b represents the structure of the motor of FIG. 2 a

FIG. 3 is a functional block circuit diagram of a control circuitembodying the present invention;

FIG. 4 is a block diagram showing a detail of the circuit of FIG. 3;

FIG. 5 a is a waveform diagram showing an ideal back EMF in a threephased motor (named a, b, and c) utilized in an embodiment of thepresent invention;

FIG. 5 b is a phase to phase back EMF derived from measuring total backEMF across two phases (a and b, b and c, a and c);

FIG. 5 c is a filtered waveform diagram of the phase to phase waveformof FIG. 5 b;

FIG. 6 is a circuit diagram of a low-pass filter that may be used in thecontrol circuit embodying the present invention;

FIG. 7 a is the binary output from three comparison operations acting onthe phase to phase back EMF signals (for example, C1=1 when Va-b>Vb-c);

FIG. 7 b is a wave form diagram of the commutation of the current to theindividual phases that may be derived by the control circuit embodyingthe present invention;

FIG. 8 a is a wave form diagram of phase currents generated when themotor of FIG. 2 b is utilised as a generator;

FIG. 8 b is a waveform diagram of the rectified phase currents shown inFIG. 8 a; and

FIG. 8 c is a waveform diagram of the rectified phase current from aconventional generator.

FIG. 1 a shows the ideal current 10 that must be supplied to each phasewinding in a prior art synchronous AC motor, as shown in FIG. 1 b, (orconversely the current generated by a prior art synchronous ACgenerator). As will be explained later, ideally a sinusoidal current 10(with commutation frequency 16) is optimal for this type of prior artmotor and care and effort is taken to provide as close a representation12 of a sinusoidal wave pattern as possible when driving such a priorart motor. One commonly used technique to achieve this ideal waveform isPulse Width Modulation (PWM). PWM involves providing a number of currentpulses 12 to the device of varying duration. By varying the averagepulse width and the timing of the pulse (switching frequency 14) anoverall current approximating a sinusoidal wave can be generated.Amplitude is varied by controlling the average pulse width andcommutation is controlled by changing the timing of the pulses.Generally, the current pulses 12 are applied with multiple phases, mostpreferentially three different phases differing by 120 degrees.

FIG. 1 b shows a prior art brushless AC motor 20 with a four polepermanent magnet rotor 22 which is mounted on a shaft 24. In this typeof motor 20, the motor has four magnetic poles spaced around itscircumference. The magnetic poles are provided by four permanent magnets26, 28, 30, 32 spread around 360 degrees; however each magnet spans only60 degrees, separated from its neighbours by 30 degrees dead space. Themagnets 26, 28, 30, 32 naturally generate a ‘blocky’north-south-north-south magnetic field around the motor.

An example of how the windings 34 of the 3-phase voltages aredistributed around the magnets as shown in FIG. 1 b. Only one loop ofthe winding 34 has been shown for clarity. It may be seen that thewinding 34 emerges from the slot 36 adjacent to magnet 28, beforepassing through the slot 36 adjacent to the edge of magnet 30. Thiswinding pattern creates differing magnetic fields within the windingsdependent upon the relative position and direction of the windingrelative to the magnets of the motor. It may, of course be appreciatedthat by varying the winding pattern the properties of the motor may betailored.

Conventional motors have their windings distributed so as to break upthe ‘blocky’, on/off excitation that would naturally be caused by themagnets. In this example of a typical motor winding pattern, 15 slots 36are available for the windings 34 so that each phase of the inputvoltage is wound around 5 slots. As 5 is not a multiple of the number ofmagnetic poles on the rotor (4) it is impossible for all 5 coils of onephase to be excited in the same way by all the magnets at the same time.Rather, the various coils of the same phase are excited by differentamounts at different times. Furthermore, the 5 coils are not evenlyspread around the stator but instead distributed as shown in FIG. 1 b.If this machine is a generator, the potential created in each coil (allof which are connected in series for one phase) changes more-or-lessindependently from one another as the rotor moves, and the distributionof the coils is chosen such that the total potential rises and falls ina nearly-sinusoidal pattern In conventional brushless DC control, sixIGBTs (A+, A−, B+, B−, C+, C−) such as illustrated in FIG. 4 areutilized to control both the commutation (timing) and voltage regulation(quantity). Voltage regulation is implemented by Pulse Width Modulation(PWM) as shown in FIG. 1 c. Pulses 12 a of constant voltage amplitudeare supplied to the motor and form a square-wave voltage 10 a. In thiscase the amplitude of the voltage 10 a is determined by the number ofpulses and their duration or width (the duty cycle). The inductance andresistance of the motor provides an inherent regulation of current.However, the inductance and resistance of the motor reduce efficiencyand make the motor unsuitable for very high electrical/mechanical speedapplications.

Additionally, to produce a relatively steady, low current from theon-off switching of the PWM controller requires the windings 34 of theDC brushless motor to exhibit high inductance. Furthermore, to implementthis sort of PWM control on the same IGBTs that control commutationmeans that the switching frequency 14 of the IGBTs must be significantlyhigher than the switching frequency of commutation 16 and higher stillcompared to the speed of rotation of the motor shaft. This makes themethod impractical at high electrical speeds. For example, IGBTswitching frequency in a PWM motor controller would typically be atleast 10× higher than the commutation frequency for a motor withsufficiently high inductance to smooth the resulting PWM output.Furthermore, in the type of motor suggested in the preferred embodimentof the present invention, which entails very low inductance and highefficiency, the IGBT switching frequency 14 would need to be at least100× higher than the commutation frequency 16. With the high operatingspeeds desired of embodiments of the present invention, this controlapproach becomes impractical.

Conversely, the motor 40 employed by the current invention uses a 12slot design. A representation of this motor is shown in FIG. 2 b. Inthis motor, the four magnets 41-44 span the full 360 degrees of therotor 46 without any dead space, creating a continuous permanent magnetshell, so the motor is (in general) 50% more powerful for a given sizecompared to the 15-slot motor. 12 slots 48 and three phases 50 allow 4coils or slots per phase, which corresponds to the 4 magnetic poles onthe rotor. Therefore each coil 50 can always be fully excited by themagnets 41-44. The coils 50 in any one phase are woundclockwise-anticlockwise-clockwise-anticlockwise, so thenorth-south-north-south magnetic field reinforces and drives the maximumcurrent through the stator (in the case of a generator) or createsmaximum torque from a given current (in the case of a motor). However,as a generator this machine would supply a square-wave output that isdifficult to deal with. Similarly as a motor 40, smooth rotation of theshaft 47 requires a square-wave current input 60 which is difficult tosupply. For these reasons, a motor with the features of this 12-slotmachine described here is not a popular choice in most prior-artapplications. If this motor were chosen for its compactness andefficiency and then driven using a prior-art PWM controller, the resultwould be unsmooth (varying with time) motor output and additionalelectrical losses, negating some of motor's inherent benefits.

FIG. 2 a shows the current 60 that must ideally be imparted upon eachphase winding 50 for a motor 40 (FIG. 2 b) designed for square-waveinput. In addition, gaps 66 are necessary as a function of statorgeometry and to prevent non-ideal excitation of the rotor when themagnetic poles of the rotor are not aligned with the permanent magnetenergised by the coil. During these gaps 66, current is imparted by adifferent phase. The relevant switching points 68 between theapplication or removal of current is the commutation timing and ideallyoccurs when the pole of the rotor passes into or out of the influence ofthe magnet energised by the windings.

In order to achieve high speed and high efficiency, the resistance andinductance of the windings 50 are much smaller than in a typical DCbrushless motor, such as the prior art motor of FIG. 1 b. At any momentduring the motor's operation, one phase 50 is connected to the positive(current travelling in), one phase is connected to the negative (currenttravelling out), and one phase is floating (no current). In order tomaximize the performance of the motor, the current should be injected toeach phase 50 when that phase exhibits the maximum back EMF compared tothe other phases and should return from each phase when that phaseexhibits the minimum back EMF. The commutation timing 68 must beaccurately controlled. If the back EMF is ideal, the commutation timing68 can be obtained by comparing the three phase voltage (e.g., when aphase exhibits the greatest back EMF, then that phase current isswitched ‘on’).

Ideally, the amplitude 62 needs to be independently variable withrespect to the commutation frequency 64. FIG. 3 shows a main embodimentof the present invention and details the controller 80 used. A principlefeature of this controller 80 is that it addresses power separately fromcommutation. This control approach is achieved by a logical separationbetween the control of aggregate current i1 82 flowing to the motor 84and the commutation of that current iu, iv, iw 86 a-c on the phaseconnectors of the motor 84.

The aggregate current 82 has two proportional-integral (PI) feedbackcontrol loops 88, 90 that regulate aggregate current 82. The inner loop88 controls the current amplitude directly and the outer loop 90 adjuststhe current in response to the torque requirement (speed/target speedmis-match) of the motor 84.

The inner loop 88 comprises a duty cycle 92 that provides the amplitudeof the aggregate current 82 and a (amplitude) regulator 94 that comparesthe present aggregate current 82 to the current requested by the outerloop 90. If the aggregate current 82 requested by the outer loop 90 isgreater than the currently supplied aggregate current then the currentis adjusted to match the desired current by the duty cycle 92. It can beappreciated that the inner loop 88 can be considered to be a regulatingfeedback loop for regulating the current amplitude.

The outer loop 90 also comprises a (speed) regulator 94 that compares aspeed target 96 with the current speed of the motor 98 and determinesthe aggregate current 82 required to accelerate to the speed target 96.A saturation check 100 is provided to ensure that the currentrequirements are within the capability of the controller 80 and themotor 84. The speed of the motor is provided by a FN converter 102 thatanalyses back EMF signals Vw, Vv, Vu 104 obtained from the motor andconverts them to determine the motor speed 98 and the angular positionof the motor (and the magnets). The components used to regulate theaggregate current 82 (the inner and outer feedback control loops 88, 90)may be considered as a current supply feedback loop for providing acurrent amplitude to the motor 84 windings.

The use of the back-EMF signal generated by the strong permanent magnetsmoving past the windings in the motor is advantageous because theback-EMF manifests itself as an oscillating variation in the apparentelectrical resistance across each phase connection of the motor's statorwinding. This gives an indication of the instantaneous position of therotor relative to the stator and thus the appropriate timing for theelectrical excitation of the stator. By this method, the motor's phaseconnections carry the output of the motor controller (oscillatingcurrent to excite the motor's stator windings) as well as one of itsinputs (back-EMF to determine the commutation pattern).

Although the present invention utilises back-EMF signals to determinemotor speed and position, alternative ways of monitoring the motor andproducing reference signals may be utilised. Examples of alternativesinclude: the use of an external rotor position sensor, most likely anoptical type or electromagnetic interference (Hall effect) sensor typeresponding to markings or shapes (e.g., compressor blades) on the motorshaft; the use of a timekeeping device internal to the controller, whichis regularly calibrated or reset (e.g., once per motor shaft rotation)by a coarse sensor; a measurement of the commutation current, or asignal indicative thereof, relating to the current induced in the motorwindings (not the total current going to the motor); and the use ofpurely internal logic and timekeeping which makes assumptions about theposition of the rotor and the required commutation without expecting orwithout caring that this may fall out of synchronicity with the true,optimal commutation timing (e.g., the rotor may ‘slip’ relative to theelectrical excitation).

This two-tier approach is implemented in order to prevent anover-current condition, because the motor 84 is optimally designed forvery low internal inductance and is therefore highly sensitive to damageunless current 82 is tightly controlled on a short timescale. To controlspeed 96, the control system 80 measures the frequency of the motor backEMF 104 to get the motor speed 98. By setting the current command 90 tothe inner loop 88, the control system can control the torque. If themotor 86 needs to accelerate, the controller 90 will increase thecurrent command to increase the torque.

The commutation of the aggregate current 82 is implemented separatelyand is shown to the right of the motor 86. The commutation pattern 110responds passively to the motor position as measured by tracking theback-EMF 104 displayed on the phase connectors. The preferred embodimentuses the phase-to-phase voltage to measure back-EMF. This would normallylead in phase by 90 degrees relative to the optimal current commutationtiming, based on the typical properties of motors (see below). Thepreferred embodiment therefore implements a low-pass filter 112 whichproduces a 90 degree phase shift in the measured phase-to-phasevoltages. This low-pass filter 112 additionally removes errors from theback-EMF signal 104 and simultaneously adjusts the phase angle so thatthe timing is appropriate for use as a current commutation controlsignal.

Once the commutation pattern 110 is determined, it is provided to theIGBT module 114. The aggregate current 84 can then be regulated by theIGBT module 114 in the required commutation pattern 110 to deliver therequired current iu, iv, iw 86 a-c to the motor 84. This combination ofcomponents 110, 112 and 114 act as a commutation feedback loop forcontrolling the timing and duration of excitation current supplied tothe motor windings.

FIG. 4 highlights the duty cycle 92 and the IGBT module 114. The dutycycle 92 acts as a “DC/DC current source” part and creates a nearlycontinuous current of controlled aggregate amperage 82. The duty cyclehas two IGBTs 120, 122 and by switching on and off the IGBTs, theaggregate current 82 can be regulated. The duty cycle 92 is connected tothe IGBT module 114, which acts for a three phase signal as a six-leginverter. Because of the high fundamental frequency of the motor, thisIGBT module 114 only controls the commutation, and need never interruptthe aggregate flow of current to control power (as it would have to doin a more conventional control layout). The “inverter” part takes asinput a commutation signal from a digital controller (not shown) and theaggregate current 82 produced by the duty cycle 92.

As output, the IGBT module 114 produces square wave current signals todrive the PM motor. The function of the IGBT module 114 is to deliverwhatever aggregate current 82 is available from the duty cycle 92directly to the motor 84 using the simple switching pattern shown inFIG. 2 a. For each phase of current 86 a-c, two IGBT's are provided. Thecommutation pattern for current iu 86 a is provided by IGBT's 116 a, 116b that switch on and off the aggregate current 82 supply to the requiredcommutation pattern 110. Similar IGBT's 118 a, 118 b, 120 a, 120 bperform the same function for each additional phase of current iv 86 b,iw 86 c. Therefore the current supplied by each phase can be eitherpositive, negative or zero.

The primary advantage of this approach is that it removes the need forIGBTs to operate at impractically high frequencies. It also enables themotor 86 to be built with lower inductance. Finally, this approachremoves from the motor's phase windings the disturbances that arenormally associated with PWM control. This makes the back EMF signals104 clearer and improves the accuracy of commutation timing 110. At highelectrical speeds, the efficiency of the controller 80 is highlysensitive to the commutation timing 110. Therefore, the additionalfeature of removing disturbances from the phase windings furtherimproves the efficiency of this approach.

The back-EMF signals 104 generated by the motor 86 are shown if FIG. 5a. The three back-EMF signals 104 a, 104 b and 104 c correspond to thethree phases of the input currents 86 a, 86 b, 86 c. The back-EMF shownin FIG. 5 a is idealised. In reality the back-EMF signal 104 is oftenchoppy and distorted, making determining the angular position of themotor and therefore determination of the commutation timing difficult.Furthermore, in practical motor control, the commutation itself disturbsthe back-EMF 104 because of the rapid phase current changes. Thisdisturbance can deform the shapes of the back-EMF wave forms, whichmakes the comparison between them no longer reliable. Additionally, dueto the practicalities of wiring the controller, measuring a singleback-EMF 104 a is difficult.

In the present embodiment, the reliability of the back-EMF signal isfurther improved by measuring the phase-to-phase voltages 130 of theback EMF 104 (allowing the controller to monitor the same wires that areused to impart current to the motor) as shown in FIG. 5 b. However, thephase-to-phase voltages 130 (back EMF's) are not aligned with the phasevoltages (back EMF's) 104. For example, the Phase A and Phase B crossingpoint (marked as Point 1) in FIG. 5 a would be the Phase A-B zerocrossing point (marked as Point 2) in FIG. 5 b. As the phase to phasecrossing point determines the optimal position for switching the currentsupplied to the motor to the next corresponding phase, determination ofthis position is critical to ensure efficient use of the motor. Thephase-to-phase voltage is the difference between two phase voltages, andso the difference in phase between these two signals can be calculatedas follows:

Phase A voltage: sin(x)

Phase B voltage: sin(x−pi/3) (120 degree phase offset in 3-phase motor)

Phase A-to-B: sin(x)−sin(x−pi/3)=sqrt(3)sin(x+pi/6) (a waveform 30degrees ahead of sin(x))

The phase to phase crossing point (point 1 in FIG. 5 a) is no longerwithin an easily determined position (point 2 in FIG. 5 b). In order toget a reliable signal, the three phase back EMF is therefore filteredbefore the comparisons are carried out. The low pass filter 112 designis shown in FIG. 6. The transfer function for the filter 112 is

$\frac{50}{{0.01S} + 1}.$

With increasing motor speed and electrical frequency, the behaviour ofthis filter 112 approaches a pure integral, and the time lag produced bythis filter approaches 90 degrees lag. A representation of the filteredphase to phase signal 140 is shown in FIG. 5 c.

Table 1 shows the degree of phase shift imparted by the filter at thedifferent speeds of the motor. It can be seen that for a large range ofthe motor speed (from 200 Hz to 2000 Hz), the phase shifts are veryclose to 90 degree.

TABLE 1 Phase shift VS RPM Motor speed Phase shift  1000 RPM  64.5degree  5000 RPM  84.5 degree 10000 RPM 87.27 degree 40000 RPM 89.31degree 120000 RPM  89.77 degree

As stated above, the ideal switching timing is obtained by consideringthe crossing points between phase voltage signals. But the controlleruses filtered phase-to-phase voltage signals, which are in total 60degrees behind the phase voltages (30 degrees-90 degrees). Because acommutation event occurs every 60 degrees (see FIG. 5 c), these filteredphase-to-phase voltages can be used, although the mapping of whichcrossover points are associated with which phase current signals isdifferent to the mapping that would apply if the phase voltages wereused.

FIG. 5 c shows the filtered phase to phase signals with 90 degree phaseshift considered. Aligned with Point 1 and 2 in FIGS. 5 a and 5 b, thecorresponding point in FIG. 5 c is marked as Point 3 which is thecrossing point between Phase B-C and Phase C-A. FIG. 5 c shows that,despite the constant phase shift, the commutation timing 110 for currentswitching can still be determined by comparing the magnitudes of thefiltered, phase-to-phase voltages.

The three voltage signals produced by the phase-to-phase voltage filtercan then be compared using well-known electronic components. The resultsof the comparisons can be decoded to generate the commutation output, ascan be seen in FIG. 7 a. C1 152 is the comparison results between thefiltered Va-b and Vb-c. C2 154 is for Vb-c and Vc-a. C3 156 is for Vc-aand Va-b. The six IGBTs 116 a, 116 b, 118 a, 118 b, 120 a, 120 b whichcontrol the commutation pattern of current to the motor can becontrolled completely and optimally by the signals C1, C2, C3, as shownin the bottom graph of FIG. 7 b. When the A+ IGBT 116 a is switched on,the positive current is injected into Phase A 166 a. When, A− 166 b ison, negative current is injected into phase A 166 b. By comparisonbetween the waveform C1 (152) of FIG. 7 a and the A+ and A− waveforms ofFIG. 7 b, it may be seen that the point when C1 152 switches fromnegative to positive occurs at π/6. This point of π/6 corresponds to thepoint at which IGBT 116 a must be switched to provide A+ current 116 a.Similarly, the point at which C1 goes from positive to negative 7π/6corresponds to switching point of A− 166 b by switch 116 b.

Thus the controller 80 imparts upon the motor's phase connections 86 a-coscillating current signals whose waveforms are shaped and spread (phaseoffset) in such a way that the sum of their absolute values is alwaysequal to the constant (aggregate current 82) signal from which they wereconstituted. This multiphase commutation pattern 110 is the controller'soutput and is sent to the phase connection points available in themotor's stator winding. This controller 80, including the commutationpart, is electronic, rather than mechanical, which improves efficiencyand reliability compared to rubbing or sliding mechanical switchesembedded in the motor and potentially moving at high speed.

It may also be appreciated that the controller 80 may be used to run themotor 84 as a generator. In such embodiments, movement of the rotor ofthe generator relative to the stator causes a current to flow withinwindings of the stator. The commutation circuit in such embodimentspulls the current off the windings, creating (in the above example) athree phase power signal. When the controller 80 operates the motor 84as a generator, the controller 80 continues to operate in the samemanner as described above, independently of the current source. However,when the motor 84 is run as a generator, the current source isessentially reversed and so the direction of the current flow is alsoreversed causing current to flow out of the commutation circuit andmotor 84. Due to the arrangement of the motor or generator, such outputis a DC signal or current. By varying this DC current between positiveand negative setpoints, the controller allows rapid and seamlesstransition from motor to generator to motor.

When run as a generator, said power signal can then be passed through arectifier to create a DC current. Advantageously, the commutationpattern produced by the generator embodiment of the present invention isa series of square waveforms as shown in FIG. 8 a and similar to thoseshown in FIG. 7 a. As may be seen from FIG. 8 a, the generator (of theform of the motor 40) produces three phase current signals 180(comparison between A and B of FIG. 7 b), 182 (comparison between B andC of FIG. 7 b) and 184 (comparison between C and A of FIG. 7 b). Each ofthe three phase signals generated 180, 182, 184, produce a signal thatvaries from a positive phase current 180 a to a negative phase current180 e via zero net phase current plateaus 108 c, 180 g and steppedfunctions 180 b, 180 d, 180 f, 180 h between the positive, negative andnet zero plateaus. It can be appreciated that the exact form of thewaveforms is likely to vary from this idealised normal representation.

The type of generator described here produces square wave output 180,182, 184 from the individual generator phases, which, when rectified toDC, is smooth 190 (except for harmonics 192). A representation of therectified DC current is shown in FIG. 8 b. Harmonics 192 may occurduring the transition from one square-wave phase output 180 to the next182 (at each step function interval). However, after harmonics 192 areremoved, the overall signal 190 is smoother than a rectified 3-phasesinusoidal signal 194 as shown in FIG. 8 c. (The signal 190 varies byapproximately 0%, rather than the 10% typical of rectified 3-phasesinusoidal signals 194.) This is desirable, since many industrialapplications are naturally capable of filtering harmonics but incapableof tolerating bulk fluctuations caused by rectifying sinusoidal signals.Furthermore, the skilled reader will appreciate that several methodsexist to filter short-duration harmonics of the type produced whenrectifying square-wave input.

If the machine is designed purely for generator operation, then thecontroller could be simplified to a rectifier, rather than the fullcontroller described here. Whatever the control method used for thegenerator, the particular combination of features described here willcreate a generator that lends itself to produce a DC power outputsignal.

1. A control system for an electric motor comprising a supply forsupplying excitation current to different windings of the motor at anygiven time, wherein the amplitude of the excitation current isindependently variable of the timing and duration of the application ofthe excitation current to the windings.
 2. A control system according toclaim 1, wherein the supply comprises a current supply feedback loop forproviding a current amplitude to the motor windings, and a commutationfeedback loop separate from the current supply feedback loop forcontrolling the timing and duration of excitation current supplied tothe motor windings.
 3. A control system according to claim 2, whereinthe current supply feedback loop further comprises a regulating feedbackloop for regulating the current amplitude.
 4. A control system accordingto claim 1, further comprising a signal provider for providing a signalindicative of the motor speed and/or the angular position of the motor.5. A control system according to claim 2, wherein the current supplyfeedback loop comprises an input for representing a target motor speed,and an output for providing a current amplitude responsive to the targetspeed of the motor and the motor speed signal.
 6. A control systemaccording to claim 4, wherein the commutation feedback loop is operativefor the control of the timing and duration of the excitation current tothe motor windings in dependence upon the angular position signal.
 7. Acontrol system according to claim 4, wherein the signal indicative ofthe motor speed and/or angular position of the motor and measured by thecontrol means is a commutation signal indicative of the current inducedin the motor windings.
 8. A control system according to claim 6, whereinthe rotation of the motor in response to the supply of excitationcurrent to any given motor winding of a plurality of phase windingsgenerates a back EMF in each phase winding that corresponds to saidsignal.
 9. A control system according to claim 8, wherein thecommutation feedback loop provides a current pulse to said given phasewinding when the amplitude of the sensed back EMF for the correspondingphase winding is greater than the back EMF sensed in respect of theother ones of the plurality of phase windings.
 10. A control systemaccording to claim 9, wherein the commutation feedback loop comprises afilter for filtering the sensed back EMF to facilitate synchronisationbetween the supply of the current to said given winding and the angularposition signal.
 11. A control system according to claim 10, wherein thefilter introduces a phase shift to the sensed back EMF to facilitatesaid synchronisation.
 12. A control system according to claim 11,wherein the said motor comprises three phase windings at 120 degreerelative angular displacement.
 13. A control system for an electricmotor, the control system comprising: a commutation circuit operativefor controlling a timing and duration of current supplied to theelectric motor; and a power supply for supplying current to the motor,wherein the commutation circuit is operationally independent of thepower supply.
 14. A permanent magnet motor comprising a control systemcomprising a supply for supplying excitation current to differentwindings of the motor at any given time, wherein the amplitude of theexcitation current is independently variable of the timing and durationof the application of the excitation current to the windings, whereinthe motor includes a plurality of permanent magnets in contact with oneanother without spaces between the magnets such as to provide acontinuous permanent magnet shell.
 15. A permanent magnet motoraccording to claim 14, wherein the permanent magnet motor furthercomprises of a plurality of phase windings disposed in a number of slotsaround the circumference of the motor such that each phase winding maybe complementary energized by a single excitation current and do notserve to counteract one another.
 16. A permanent magnet motor accordingto claim 15, wherein the plurality of phase windings are individuallydisposed within the slots in parallel.
 17. A permanent magnet motoraccording to claim 15, wherein the number of slots divided by the numberof magnets is a multiple of the number of phases of the current. 18.(canceled)
 19. A forced induction system for an internal combustionengine with a crankshaft, the system comprising: a compressor forincreasing the pressure of gas into the engine; a turbine mechanicallydecoupled from the compressor and arranged to be driven by engineexhaust gas; a generator arranged to be driven by the turbine; anelectric motor arranged to drive the compressor, wherein the generatorand motor are electrically connected; and a control system comprising asupply for supplying excitation current to different windings of themotor at any given time, wherein the amplitude of the excitation currentis independently variable of the timing and duration of the applicationof the excitation current to the windings whereby the compressor isdriven at least in part by an output torque of the turbine via theelectrical connection.
 20. A permanent magnet generator including aplurality of permanent magnets in contact with one another withoutspaces between the magnets such as to provide a continuous permanentmagnet shell.
 21. A permanent magnet generator according to claim 20,wherein the permanent magnet generator comprises a plurality of phasewindings disposed in a number of slots around the circumference of thepermanent magnet shell such that each phase winding may becomplementarily energized by a single excitation current and do notserve to counteract one another.
 22. A permanent magnet generatoraccording to claim 21, wherein the plurality of phase windings areindividually disposed within the slots in parallel.
 23. A permanentmagnet generator according to claim 21, wherein the number of slotsdivided by the number of magnets is a multiple of the number of phasesof the current.